Apparatus and method for improved chopping mixer

ABSTRACT

An apparatus and method for an improved chopping mixer ( 100 ) having a bipolar mixer stage ( 140 ) for mixing signals (lp, In, LOp, LOn) received thereby; an output chopping stage ( 160 ); and an AC coupling stage ( 150 ) for coupling the mixed signal to the output chopping stage. The signal prior to the chopping output stage is centered at the chopping clock frequency rather than DC. AC coupling allows removal of common mode signal in a desired frequency range. Also, the second order component present on each single ended output will also be DC blocked by the AC coupling capacitors, resulting in improved second order IP2 performance.

FIELD OF THE INVENTION

This invention relates to chopping mixers, and particularly (though notexclusively) to radio frequency circuitry such as direct conversionreceivers.

BACKGROUND OF THE INVENTION

In the field of this invention it is known to use direct conversion orzero-IF receivers in radio receivers for applications such as cellulartelephony. In such receivers, it is necessary to maintain the spectralpurity of the channel used for reception. Because of limited narrow bandselectivity, second order intermodulation distortion (IM2) presents anundesired spectral component within the signal band of interest. Thisoccurs when two or more interfering signals, whose difference infrequency is less than the IF bandwidth of the desired signal, mix withone another due to some second order nonlinearity and produce a basebandspectral component. To minimize the effects of second orderintermodulation within critical circuit blocks in the signal path, it isknown in the art to use differential circuits. In theory, differentialcircuits have infinite attenuation to second order intermodulationdistortion; however, in reality this is far from the truth, due in nosmall part to device mismatches, parametric imbalance, imperfect layout,and other device characteristic inequalities that cause imbalances whichprovide a lower than desired second order input intercept point (IIP2).As will be appreciated by those skilled in the art, the best IIP2achieved to date in the integrated mixer art may fall significantlyshort of system requirements. It would be extremely advantageous,therefore, to provide improved chopping mixer performance so as to allowthe above difficulties to be overcome. It would be of greater advantageto apply this improved chopping mixer performance to wireless andwireline communications, devices that employ mixer circuits, switches,and other components that exhibit parametric mismatch or imbalance.

United States patent U.S. Pat. No. 5,859,559 (RAYTHEON) describes amixer structure suitable for inclusion as part of an integrated circuit.Spurious signals is avoided by introducing trickle currents whichenhance the transconductance of an input differential amplifier.

United Kingdom Patent Application No. GB-A-2 151863 (Toshiba) describesan amplifier circuit having first and second differential amplifiers. Aswitch circuit enables the dynamic range of the output signals to beincreased. First and second outputs are applied to a load so as toobtain the product of the first and second signals.

It is an object of the present invention to provide method and apparatusfor improving chopping mixer performance wherein the abovementioneddisadvantage(s) may be alleviated.

STATEMENT OF INVENTION

In accordance with a first aspect of the present invention there isprovided a chopping mixer as claimed in claim 1.

In accordance with a second aspect of the present invention there isprovided a method of operating a chopping mixer as claimed in claim 13.

BRIEF DESCRIPTION OF THE DRAWINGS

Method and apparatus for improving chopping mixer performance utilisingthe present invention will now be described, by way of example only,with reference to the accompanying drawing(s), in which:

FIG. 1 shows a schematic circuit diagram of a first AC chopping mixer ina direct conversion radio receiver;

FIG. 2 shows a waveform timing diagram of clock signals used in the ACchopping mixer of FIG. 1; and

FIG. 3 shows a schematic circuit diagram of a second AC chopping mixer.

DESCRIPTION OF PREFERRED EMBODIMENTS

Referring firstly to FIG. 1, an AC chopping mixer 100 for use in adirect conversion radio receiver 110 is shown. The mixer 100 has aninput chopper cell 120 constituted of two pairs of MOSFET choppertransistors 122 & 124 and 126 & 128. The chopper transistors 122-128 arecoupled to receive chopper clock signals clkp and clkn (see FIG. 2) andcross-coupled differential input signals RFp and RFn at an RF inputport. The input chopper cell 120 produces chopped differential voltageoutput signals Vip and Vin.

A voltage-current (V-I) converter 130 is constituted of bipolartransistors 132 and 134 that have their base electrodes commonly coupledto receive a bias voltage Vb. Outputs from the chopper transistors122-128 of the input chopper cell 120 are connected respectively toemitter electrodes of the bipolar transistors 132 and 134 in thevoltage-current converter 130, so that the transistors sink at theircollector electrodes currents Ip and In that are proportional torespectively the voltage output signals Vip and Vin from the inputchopper cell 120.

A radio frequency (RF) mixer cell 140 is constituted of two pairs ofbipolar transistors 142 & 144 and 146 & 148. The base electrodes of thetransistors 142 and 148 are commonly coupled to receive an input signalLOn, and the base electrodes of the transistors 144 and 146 are commonlycoupled to receive an input signal LOp, the signals LOp and LOn forminga differential input signal (at an LO input port) to be mixed with thedifferential input signal RFp, RFn (at the RF input port). The emitterelectrodes of the bipolar transistors 142 and 144 are commonly connectedto the collector electrode of the bipolar transistor 132, and theemitter electrodes of the bipolar transistors 146 and 148 are commonlyconnected to the collector electrode of the bipolar transistor 134, ofthe voltage-current converter 130. The collector electrodes of thebipolar transistors 142 and 146 are commonly coupled (via a resistanceRip) to a source of reference potential Vpp, to which the collectorelectrodes of the bipolar transistors 144 and 148 are also commonlycoupled (via a resistance Rin).

An AC coupling cell 150 is constituted of capacitors Cn and Cp. Oneelectrode of the capacitor Cn is connected to the commonly connectedcollector electrodes of the bipolar transistors 142 and 146, and oneelectrode of the capacitor Cp is connected to the commonly connectedcollector electrodes of the bipolar transistors 144 and 148. As will beexplained below, the capacitors Cn and Cp may be realised as aprogrammable capacitor structure (not shown) to allow their capacitanceto be varied.

An output chopper cell 160 is constituted of two pairs of MOSFET choppertransistors 162 & 164 and 166 & 168. The chopper transistors 162-168 arecoupled to receive chopper clock signals clkp and clkn and are connectedto the AC coupling cell 150 to the receive voltage signals Vnc and Vpcfrom the capacitors Cn and Cp respectively. The outputs of the chopperelements 162 and 164 are cross-coupled to produce differential outputsignals Von and Vop (at a BB output port), which are mixed from thedifferential input signals RFn & RFp (at the RF input port) and LOp &LOn (at the LO input port).

The performance of the AC chopping mixer 100 circuit may be analysed asfollows:

At the chopper output stage, the differential output versus thedifferential input relation is given by:Vop=Vpc and Von=Vnc when clkp is active (clkp=1, clkn=0)Vop=Vnc and Von=Vpc when clkn is active (clkp=0, clkn=1).This results in the equalitiesVop−Von=Vpc−Vnc when clkp is active, andVop−Von=−(Vpc−Vnc) when clkn is active.Thus, the differential input Vp−Vn is multiplied by the clock signal clksuch the signal Vp−Vn is downconverted from clk to DC, soVop−Von=(clkp−clkn)(Vpc−Vnc).However, for the common mode the situation is different, i.e., thecommon mode output versus the common mode input relation is given by:Vop+Von=Vpc+Vnc when clkp is active, andVop+Von=Vpc+Vnc when clkn is active.Thus, the common mode at the chopper stage output is the same as theinput, soVop+Von=Vpc+Vpn.Thus, it can be seen that the chopper stage does not change the commonmode signals input.

Also it will be understood that the RF mixer cell will behave like thechopper stage, i.e., the differential output versus the differentialinput relation will be given by the following:When LOn is active (LOn=1, LOp=0), thenVp−Vcc=−Rip.Ip and Vn−Vcc=−Rin.In, andWhen LOp is active (LOn=0, LOp=1), thenVp−Vcc=−RiIp and Vn−Vcc=−RinIp.So,Vp−Vn=Ri(In−Ip) when LOn is active, andVp−Vn=Ri(Ip−In) when LOp is active.Therefore, Vp−Vn=(LOp−LOn)Ri(Ip−In).Also, the common mode output versus the common mode input relation isgiven by:Vp+Vn=−Ri(Ip+In)+2 Vcc when LOn is active, andVp+Vn=−Ri(Ip+In)+2 Vcc when LOp is active,so it will be appreciated that the RF mixer stage does not change thecommon mode signal input.

Although the V-I converter is idealized as a linear stage, it will inpractice introduce non-linear behavior on the output current that couldbe modelled as a polynomial relation versus the voltage input Vip andVin. Considering only the second order output current and discarding theuseful signal gives the following relations:Ip=a2p(Vip)²,In=a2n(Vin)²

The generated second order current components will appear at lowfrequency. These second order currents generate a common mode currentsignalIp+In=a2p(Vip)² +a2n(Vin)²which will result at the RF mixer output as a common mode voltagesignal, i.e.,Vp+Vn=−Ri(a2p(Vip)² +a2n(Vin)²)+2Vccthat occupies the same spectrum as the current signal, i.e., at lowfrequency, so that the AC coupling network will reduce the content inthose low frequencies (the AC coupling corner frequency may be chosen tolie in the range of approximately 2.5% to 5% of the chopping clockfrequency, and may be variable within this range by use of aprogrammable capacitor structure as mentioned above).

Also, considering the single ended voltage at the RF mixer output forthose second order components (supposing now that Rip is different fromRin (i.e., there is resistor mismatch), produces the followingrelations:Vp=Vcc−Rip a2p (Vip)² when LOn is active, andVp=Vcc−Rip a2n (Vin)² when LOp is active,allowing Vp to be expressed as:Vp=Vcc−Rip(a2p(Vip)² +a2n(Vin)²)/2+(LOp−LOn)Rip(a2p(Vip)² −a2n(Vin)²)/2,and Vn to be expressed asVn=Vcc−Rin(a2p(Vip)² +a2n(Vin)²)/2−(LOp−LOn).Rin.(a2p(Vip)²−a2n(Vin)²)/2.On both Vp and Vn, the term Rix (a2p (Vip)²+a2n (Vin)²)/2 appears as thelow frequency common mode variation due to the second order componentsthat will be DC blocked by the capacitors and reduced. The term(LOx−LOn) Rix (a2p (Vip)²−a2n (Vin)²)/2 will be shifted by the localoscillator frequency so it will be easy filtered such that itscontribution is minimized.

Discarding the second term allows Vpc and Vnc to be expressed as:Vpc=Hpc*Vcc−Rip(a2p(Vip)² +a2n(Vin)²)/2*Hpc,

-   -   where Hpc is a high pass filter on the positive path and *        denotes a convolution operation, and        Vnc=Hnc*Vcc−Rin(a2p(Vip)² +a2n(Vin)²)/2* Hnc,    -   where Hnc is a high pass filter on the negative path and *        denotes a convolution operation.

At the output chopper, the following relations are satisfied:Vop+Von=Vpc+Vnc, andVop−Von=(clkp−clkn)(Vpc−Vnc),producing the relations:Vop+Von=(Hpc+Hnc)*Vcc−(a2p(Vip)² +a2n(Vin)²)/2*(RipHpc+RinHnc),

-   -   the second order common mode component being reduced by the AC        coupling, and        Vop−Von=(clkp−clkn)(Hpc−Hnc)*Vcc+(clkp−clkn)(a2p(Vip)²        +a2n(Vin)²)/2)*(RinHnc−RipHpc).        This assumes a perfectly matched output chopper stage, i.e.,        {clkp}={clkn}. However, a non-perfectly matched output chopper        (e.g., due to non−50% duty cycle clock or non-similar switches)        will limit the reduction of differential second order component        and will generate a term value that is equal to        (dutycycle−50%)(a2p (Vip)²+a2n(Vin)²)/2*(RinHnc−RipHpc).

In this case, the AC coupling will provide additional IP2 gainimprovement versus a non-AC-coupling network if (Rin Hic−Rip Hpc) ismuch smaller than (Rin−Rip) in the low frequency region (0 to 200 KHz).However, the second order non-linearities that are introduced by theoutput chopper will limit the IM2 differential performances.

Referring now to FIG. 3, a second AC chopping mixer 200 (which may beused as an alternative to the AC chopping mixer 100 described above)shares many components with the mixer 100, and these shared componentsare given the same reference numerals in FIG. 3 as in FIG. 1.

The second AC chopping mixer 200 differs from the mixer 100 in that(whereas in the first mixer 100 the LO input is coupled directly to themixer cell 140, and the RF input is coupled to the mixer cell 140 via achopping cell 120 and a voltage-current converter cell 130) in thesecond mixer 200 the LO input is coupled to the mixer cell 140 via achopper cell 220 (constituted of MOSFET chopper transistors 222, 224,226 and 228), and the RF input is coupled to the mixer cell 140 via thevoltage-current converter cell 130. It will be seen that in the secondmixer 200, as in the mixer 100, the output of the mixer cell 140 iscoupled via an AC coupling cell 150 to an output chopping cell 160,whose output is connected to the BB output.

It will be appreciated that the second mixer 200 functions similarly tothe mixer 100 described above. It will also be appreciated that both thefirst and second mixers will respond in a non-ideal manner to signals atthe RF input port that differ from the local oscillator frequency at theLO input port by an integer multiple of the frequency fclk of thechopping clock signals clkn and clkp. However, it will be understoodthat in the second mixer 200 this non-ideal signal response isdiminished, relative to the first mixer 100, by movement of the inputchopping stage from the RF input port in the first mixer to the LO inputport in the second mixer. In this way, it will be appreciated, thesecond mixer 200 offers better performance than the first mixer 100through higher spurious response isolation.

It will be understood that the method and apparatus for improvingchopping mixer performance described above provides the followingadvantages:

-   -   By AC coupling the bipolar mixer output when it is operating in        chopping mode, an advantage is gained since the signal prior to        the output chopping stage is centered at the chopping clock        frequency rather than at DC.    -   Also, AC coupling allows removal of the common mode signal in        the required frequency range of 0-200 KHz.    -   Also, the second order component present on each single ended        output will be also DC blocked by the coupling capacitors, which        results in an improvement in the second order IP2.    -   The chopper mixers and the RF bipolar mixer have similar        behavior, their common mode output versus common mode input is        equal to 1 and do not result on any frequency translation;        however, the differential input will be translated in frequency.        This means that only low frequency common mode signals (0-200        khz) that are generated within the whole mixer arrangement will        be at the output as low frequency common mode signals. Most of        these low frequency common mode signals are generated by the        second order non-linearities of the mixer arrangement, mainly in        the V-I converter stage, but the AC coupling will block those        components and reduce their levels.

1. A chopping mixer comprising: input chopping means for chopping aninput signal applied thereto and for producing therefrom a chopped inputvoltage signal; mixer means for mixing signals received thereby and forproducing a mixed signal therefrom; output chopping means; couplingmeans for coupling the mixed signal to the output chopping means; andvoltage to current converter means coupled between the input choppingmeans and an input of the mixer means for converting a voltage signal toa current signal for applying to the mixer means a current signalrepresentative of the chopped input voltage signal; wherein the couplingmeans comprises AC coupling means arranged to filter out second orderdistortion in the signal from the mixer means.
 2. The chopping mixer ofclaim 1 wherein the AC coupling means comprises capacitance means forpassing AC signals while blocking DC signals.
 3. The chopping mixer ofclaim 1 wherein the mixer means comprises a bipolar transistor mixer. 4.The chopping mixer of claim 1 wherein the output chopping meanscomprises MOSFET transistors.
 5. The chopping mixer of claim 1 furthercomprising input chopping means for chopping an input signal appliedthereto and for coupling the chopped input signal to the mixer means. 6.The chopping mixer of claim 5 wherein the input chopping means comprisesMOSFET transistors.
 7. The chopping mixer of claim 1 wherein the signalsare differential signals.
 8. The chopping mixer of claim 1 wherein theAC coupling means has a corner frequency that is in the range ofapproximately 2.5% to 5% of the frequency of the chopping means.
 9. Adirect conversion radio receiver comprising a chopping mixer as claimedin claim 1.